1. Field of the Invention
The present invention relates to improving communication signal amplification efficiency.
2. Background
Power amplifiers are more efficient when operated close to their maximum power output rating. For example, a 100 Watt power amplifier will be more efficient when it is outputting 90 Watts than when it is outputting half a Watt. With certain types of communication signals (e.g., frequency modulation), the amplitude of the signal is relatively constant, and it is possible to operate the amplifier near its maximum power rating at all times. But when the amplitude of the signal being amplified is sometimes small and sometimes large, this is no longer possible. For if an amplifier is designed to amplify peaks of the signal without distortion, the amplifier will not be operating close to its maximum power output rating whenever the it is not amplifying a peak.
The peak to average ratio is a measure of how far the peak power of a signal exceeds the average power of that signal. When a signal with a large peak to average ratio is amplified, the amplifier will spend most of its time operating far away from its maximum power output rating, and will be less efficient. Unfortunately, many modern communication signals have large peak to average ratios, so their amplifiers operate with very low efficiencies. Spread-spectrum signals (such as CDMA) are particularly bad offenders, with typical peak to average ratios ranging from six to thirteen dB.
One way to increase the operating efficiency of an amplifier is by modifying the signal to reduce its peak to average ratio before it is amplified. Of course, any modification to a communication signal will corrupt or distort the signal that is ultimately received at its end destination. But in most applications, some level of signal distortion is usually tolerable.
One prior art approach for reducing the peak to average ratio of an input waveform is to implement power clipping. In the power clipping approach, whenever the amplitude of the input signal is lower than a predetermined threshold, the input signal is passed to the output unchanged, and whenever the amplitude of the input signal exceeds the threshold, the output signal is clamped to the threshold level. Of course, the clipping operation destroys some of the information contained in the original signal. But the user should be able to tolerate this loss of information as long as the threshold is kept sufficiently high.
If an ideal power clipper with an infinite bandwidth were available, and the input signal 710 shown in FIG. 7A were applied to that ideal power clipper, the resulting output signal would resemble the curve 720 as shown in FIG. 7B. In real-world situations, however, deviations from this ideal behavior arise due in part to the limited bandwidth that may be transmitted into real-world channels.
FIG. 1 is a block diagram of a practical prior art approach for implementing power clipping. Signals arriving at input node 105 contain in-phase and quadrature (I and Q) components. These I and Q components are extracted in any conventional manner (not shown) and converted into amplitude and phase components (A and Φ) by the rectangular-to-polar converter 110. The amplitude component A is then provided as an input to an amplitude clipper 115. In addition, a threshold Th (generated by block 120) is provided to the amplitude clipper 115. Whenever the amplitude of the input signal A is lower than the threshold Th, the amplitude clipper 115 generates an output A′ that is equal to the input amplitude. Whenever the amplitude A exceeds the threshold Th, the amplitude clipper 115 generates an output A′ that is equal to the threshold value Th.
The clipped amplitude signal A′ and the original phase signal are then provided to a polar-to-rectangular converter 125, which generates corresponding I and Q components. These components are then combined in any conventional manner (not shown) and provided to the up sampler 130 which increases the sample rate of the signal from the chip rate (C×1) to twice the chip rate (C×2). The output of the up sampler 130 is then filtered by a baseband filter 135, and the output 140 of the baseband filter 135 is provided to the succeeding stage of circuitry. The baseband filter 135 is necessary because output signals in real-world systems are only permitting to occupy a finite bandwidth.
Unfortunately, when signals with high-frequency components (like the waveform 720 shown in FIG. 7B) are processed by a practical baseband filter 135, the output 140 of the filter 135 will usually overshoot the threshold level Th. This overshooting undermines the effectiveness of the clipping function. To insure that the output signal always stays below the threshold level Th when a practical baseband filter 135 is used, the threshold generated by block 120 must be reduced to some level Th′ that is below Th. Reducing the threshold to Th′, however, destroys additional information, which may not be acceptable to the user. For example, for CDMA communication signals, it is difficult to achieve a peak to average ratio of less then 8 dB using power clipping without experiencing unacceptable levels of signal information destruction.
Decresting is a second prior art approach for reducing the peak to average ratio of an input waveform, while avoiding the overshoot problems caused by the baseband filter in the power clipper. In this approach, an error signal is created that represents the amount by which the input signal exceeds a threshold. This error signal is then subtracted from the original input signal in order to form a decrested output signal.
FIG. 2 is a block diagram of a prior art system for implementing decresting. The circuit shown in FIG. 2 is designed to operate after the baseband filter at a sample rate of two times the chip rate (C×2). Signals arriving at input node 205 contain I and Q components which are extracted and converted into amplitude and phase components (A and Φ) by the rectangular-to-polar converter 210. The amplitude component A is then provided as an input to a crest detector 220. In addition, a threshold Th (generated by block 215) is provided to the crest detector 220. Whenever the amplitude of the input signal A is lower than the threshold Th, the crest detector 220 generates an output E of zero. Whenever the amplitude A exceeds the threshold Th, the crest detector 220 generates an output E of A-Th. For example, if the input signal 730 shown in FIG. 7C is applied to the input node 205, the output E of the crest detector 220 would look like the wave 740 shown in FIG. 7D. Note that the signals processed by the circuits described herein are actually sampled versions of the analog waveforms shown in FIGS. 7A-7H.
The output E of the crest detector 220 and the original phase signal Φ are then provided to a polar-to-rectangular converter 225, which generates corresponding I and Q components. The output of the polar-to-rectangular converter 225 is filtered by the error filter 230. This error filter 230 is typically implemented as a finite-impulse-response (FIR) filter, and its characteristics are selected to meet out of band emission specification (because signals in real-world systems are only permitting to occupy a finite bandwidth). The filtered output of the error filter 230 is then subtracted from a delayed version of the input signal that has been generated by the delay element 240. Because only the error signal is filtered, the filtering process can only introduce overshoot into the error signal. And because the filtered error signal is subtracted from the original input signal, any overshoot will result in an output that is smaller than the threshold (rather than larger, as with the power clipper). Therefore, any overshoot introduced by the error filter does not increase the peak to average ratio of the signal being processed.
The prior art decresting arrangement of FIG. 2 works well for the error signal 740 shown in FIG. 7E, where the crest in the error signal 740 only contains a single sample 745, and when that sample 745 coincides with the peak of the crest. But when a crest in the error signal contains more than one sample, and/or when a crest in the error signal contains one sample that does not coincide with the maximum of the crest, the accuracy of the FIG. 2 decresting system is not as good. This accuracy problem is primarily caused by the impulse response of the error filter 230.
FIG. 7F is an example of one type of problematic input signal 750 with a crest that remains above the threshold Th for so long that two or more samples will occur during the crest. When this input signal 750 is provided to the decresting system of FIG. 2, the output of the crest detector 220 will look like the error signal 760 shown in FIG. 7G. Because this error signal 760 endures for such a long time, it will contain two samples 765A and 765B, as shown in FIG. 7H. When these two samples are processed by the FIR error filter 230 in quick succession, the second sample 765B will start to contribute to the output of the FIR filter before the contribution of the first sample 765A to the filter's output has had a chance to settle down (i.e., to decay). When this happens, the resulting output of the FIR error filter 230 will be the sum of the responses to each of the two input pulses 765A, 765B. This causes the output of the error filter 230 to be too large (i.e., to rise even higher than the amplitude of the error signal 760). When this too-large signal is subtracted from the delayed version of the input signal, the resulting output at node 245 will be lower then the threshold Th. This overcompensation undesirably increases the distortion of the output signal.
Another example of a problematic input signal is when a sample does not coincide with the maximum of the crest. In this situation, the amplitude of the sample that is provided to the input of the FIR error filter 230 would be smaller than the amplitude of the crest in the input signal. As a result, the error signal E generated by the error filter 230 would also be too small. When this too-small error signal is subtracted from the original input signal, the amplitude of the resulting output signal at node 245 ends up being larger than the threshold Th, so that the desired peak to average ratio is not attained. In order to avoid this undesirable result, the threshold Th may have to be adjusted to a setting Th′ that is lower than Th. This lower threshold Th′, however, may result in a loss of signal quality that is unacceptable to the user.
The inventors have recognized a need to reduce the peak to average ratio of signals while minimizing the loss of signal quality, in order to improve the amplification efficiency of those signals.